Cross-talk cancellation scheme for rll-based storage systems

ABSTRACT

The invention relates to Run length Limited-codes storage systems. In modern storage systems, the inter-track spacing is chosen to be relatively small to allow for high storage densities. As a result, when reading a target track, data written on side tracks may appear in the recovered signal. This inter-track interference is called cross-talk. The invention proposes a cross-talk cancellation scheme based on the minimization of the mismatch between the actual (dm+1m) and the expected (exp) run length between two transitions (xm, xm+1) of the (dm+1,m) signal. The proposed solution significantly improves the ramp-up properties of the receiver and allows more efficient hardware implementation.

FIELD OF THE INVENTION

The present invention relates to a cross-talk cancellation method, acomputer program for implementing a cross-talk cancellation method, asignal processor comprising cross-talk cancellation means, and anapparatus for reading a signal stored along a track on a storage medium,said apparatus comprising cross-talk cancellation means.

The present invention relates to storage systems in which data arestored along tracks on a storage medium. In modern storage systems, theinter-track spacing is chosen relatively small to allow for high storagedensities. As a result, when reading a target track, data written onside tracks may appear in the recovered signal. This inter-trackinterference is called cross-talk.

The invention is advantageously used in such storage systems to improvethe recovered signal by removing the cross-talk. For example, theinvention applies to optical storage systems (DVD, Blu-ray Disc, SmallForm Factor Optical Disc . . . ), magnetic storage systems (hard disksnotably), magneto-optical storage systems.

With optical storage systems, the cross-talk is even more severe whenradial tilt is present in the system because then the optical spotextends more onto the side tracks.

BACKGROUND OF THE INVENTION

A cross-talk removing device is described in U.S. Pat. No. 6,134,211.This device has three reading elements simultaneously reading a maintrack and two adjacent tracks. The three signals that are read by thethree reading elements are sampled so as to provide three sequences ofsamples. A cross-talk removing circuit applies adaptive signalprocessing (for example an LMS adaptive algorithm) to the threesequences of samples to produce a cross-talk-removed sequence of samplesassociated with the main track that is free of cross-talk componentsfrom the adjacent tracks.

The adaptive processing comprises an adaptive filtering, the filtercoefficients being updated so as to converge to zero an error valuepresent in the cross-talk-removed sequence of samples.

This convergence is achieved by using a reference sample extractingcircuit. When the values of three successive samples transit frompositive to negative or from negative to positive, the reference sampleextracting circuit extracts the central sample value of three successivesample values. The extracted sample value is supplied to a subtractorthat calculates the difference between the extracted sample value and areference value. This difference is used as the error (e) that has to beconverged to zero to update the filter coefficients.

In this scheme it is assumed that the central sample value is the samplevalue at ideal zero-crossing time. This assumption can only be made ifthe samples are bit-synchronous samples.

In U.S. Pat. No. 6,134,211 this is achieved by running theanalog-to-digital converters and the cross-talk removing circuit on aclock that is driven by a time recovery circuit. As a consequence, thecross-talk cancelling scheme is only operational when the time recoverycircuit has acquired both the frequency and the phase lock.

In U.S. Pat. No. 6,134,211, when the sample sequences remain in anasynchronous state (that is, when the time recovery circuit is notlocked), the sequences are filtered on the basis of fixed predeterminedcoefficients. This helps to avoid divergence of the filter coefficientsbut leads to a ramp-up problem: if the time recovery circuit cannotconverge because of strong cross-talk, the cross-talk cancellationscheme will remain inefficient and the system will be stuck.

Another problem of the prior art system is that it is hardly compatiblewith asynchronous receiver architectures.

In such asynchronous architectures, the analog-to-digital converters andthe filters are run on a fixed clock. A transition from the fixed clockdomain to the bit-synchronous domain is done at the output of thecross-talk cancellation circuit by means of a sample rate convertercontrolled by a time recovery circuit. An additional sample ratecontroller, locked to the time recovery circuit, would be needed foreach adjacent track to produce the bit-synchronous samples that areneeded to derive the above-described error (e).

Moreover, if the fixed clock (at which the filters are running) and theclock driven by the time recovery circuit (at which the filterscoefficients are updated) differ substantially from each other, inversesample rate converters would also be required to interpolate the filtercoefficients from the domain of the clock driven by the time recoverycircuit to the domain of the fixed clock.

This would lead to an increased complexity of the architecture.

SUMMARY OF THE INVENTION

One of the objects of the invention is to propose a solution forcross-talk cancellation that solves the above-mentioned problems.

This is achieved with a cross-talk cancellation method as claimed inclaim 1, a program as claimed in claim 2, a signal processor comprisingcross-talk cancellation means as claimed in claims 3 to 5, and anapparatus for reading a signal stored along a track on a storage mediumas claimed in claims 6 to 8.

The cross-talk cancellation means according to the invention areintended for receiving a main signal associated with a target track andsatellite signals associated with side tracks, said main signal showingtransitions and runs of various lengths between two transitions. Theycomprise:

filtering means for filtering the satellite signals with adaptivefilters, thereby generating filtered versions of the satellite signals,

updating means for updating the coefficients of the adaptive filters byminimizing the mismatch between the actual and the expected run lengthbetween two transitions of the main signal,

processing means for generating an improved main signal from said mainsignal by subtraction of said filtered versions of the satellitesignals.

According to the invention the error that is to be minimized whenupdating the filter coefficients is the mismatch between the actual andthe expected run length between two transitions of the main signal.Contrary to the prior art minimization scheme, the minimization schemeof the invention does not use the notion of ideal transition time.Therefore it does not require the use of bit-synchronous samples. Theproposed minimization scheme requires frequency lock but not phase lock.

A first advantage of the proposed minimization scheme is that itresolves the above-mentioned ramp-up problem.

A second advantage of the proposed minimization scheme is that it can beimplemented in an asynchronous architecture without any additionalhardware complexity.

Advantageously, when used in an asynchronous receiver having a bit clockthat is driven by a time recovery circuit, the cross-talk cancellationmeans of the invention are operated at a fixed clock that isasynchronous with respect to this bit-clock.

In such a case, if the bit clock frequency is different from the fixedclock frequency, additional time recovery means are provided to derivethe ratio between the bit clock frequency and the fixed clock frequency,said ratio being used by said updating means for updating saidcoefficients.

BRIEF DESCRIPTION OF THE DRAWINGS

These and other aspects of the invention will be further described withreference to the following drawings:

FIGS. 1 and 2 are functional block diagrams of examples of an apparatusaccording to the invention for reading a storage medium;

FIGS. 3 and 4 are schematic representations of a first and a secondconfiguration of tracks and light spots used in a 3-spot cross-talkcancellation scheme;

FIG. 5 is a functional block diagram of the cross-talk cancellationmeans according to the invention;

FIG. 6 is a schematic representation of a received signal showing twotransitions and a run between the two transitions.

DESCRIPTION OF PREFERRED EMBODIMENTS

The invention applies to storage media having tracks each forming a 360°turn of a spiral line. Encoded data are recorded along the tracks. Theencoding scheme that is used in optical recording system is a Run LengthLimited encoding scheme (RLL). When the data recorded along the tracksare encoded with an RLL encoding scheme, the tracks exhibit markscorresponding to runs of a same value, and the edges of a markcorrespond to a transition between two runs. The size of the markcorresponds to the length of the run. It is an integer multiple of areference unit size mark.

FIGS. 1 and 2 show block diagrams of a first and a second example of anapparatus for reading such a disc. The apparatus shown in FIG. 1 carriesreference number 6-1. The apparatus of FIG. 2 carries reference number6-2. According to FIGS. 1 and 2, the apparatuses 6-1 and 6-2 comprise anoptical unit 8 having three reading elements: a main reading element 12for reading a main signal associated with a main track, and twosatellite reading elements 11 and 13 for reading two satellite signalsassociated with the two tracks that are adjacent to the main track. Inthe subsequent description, one of these satellite signal is calledupper satellite signal, and the other satellite signal is called lowersatellite signal. The three reading elements transmit three light spots21, 22 and 23.

FIGS. 3 and 4 show the locations of the three light spots 21, 22 and 23with respect to the three tracks to be read 31, 32 and 33. The mainlight spot 22 is centered on the main track 32. The satellite lightspots 21 and 23 may be centered either on the satellite tracks 31 and 33as represented in FIG. 3, or between the main track 32 and the adjacenttracks 31 and 33 as represented in FIG. 4. The satellite signals read bythe satellite light spots 21 and 23 in FIGS. 3 and 4 are said to be“associated with” the adjacent tracks because the light spots 21 and 23overlap with at least part of the adjacent tracks.

The embodiment of FIG. 4 is advantageous for rewritable optical discsystems because it allows reusing the 3-spot push-pull radial trackingmeans which are currently available in all such systems (the signal readby the reading elements 11, 12 and 13 and the main light spots 21, 22and 23 can be used both for tracking and for cross-talk cancellation).

Returning to FIGS. 1 and 2, the three signals that are read by the threereading elements 11, 12 and 13 are input to a signal processor 40comprising cross-talk cancellation means 42 and decoding means 44. Thesignal produced by the decoding means 44 is input to a reproductioncircuit 46 that generates an output signal (for example an audio or avideo signal).

FIG. 5 is a functional representation of the cross-talk cancellationmeans 42. The cross-talk cancellation means 42 comprise threeanalog-to-digital converters 51, 52 and 53 for sampling the main signal,the upper satellite signal and the lower satellite signal. The threeanalog-to-digital converters 51, 52 and 53 operate at a fixed clock 55and generate a sequence of main samples 62, a sequence of lowersatellite samples 61, and a sequence of upper satellite samples 63. Thesequences of lower and upper satellite samples 61 and 63 are processedby a lower adaptive filter 71 and an upper adaptive filter 73,respectively, which generate a filtered version 81 of the sequence oflower satellite samples and a filtered version 83 of the sequence ofupper satellite samples. The sequence 62 of main samples is processed byan optional equalizer 90, which generates an equalized sequence of mainsamples 92. Then a subtractor 93 subtracts the filtered version 81 ofthe sequence of lower satellite samples and a filtered version 83 of thesequence of upper satellite samples from the equalized sequence 92 ofmain samples, thereby generating an improved sequence of main samples102.

Alternatively, if the equalizer 90 is omitted, the improved sequence ofmain samples 102 is generated by subtraction of the filtered version 81of the sequence of lower satellite samples and a filtered version 83 ofthe sequence of upper satellite samples from the sequence 62 of mainsamples.

The improved sequence of main samples 102 is input to a sample rateconverter 120 driven by a time recovery circuit 130 (for example a PhaseLock Loop circuit). The output of the sample rate converter 120 is theinput of the decoding means 44.

The improved sequence of main samples 102 and the sequences of lower andupper satellite samples 61 and 63 are processed by lower and uppercoefficient updating means 111 and 113. The lower and upper coefficientupdating means 111 and 113 update the respectively coefficients used bythe lower filter 71 and by the upper filter 73.

The behaviour of the cross-talk cancellation means 42 can be formalizedby the following mathematical expression: $\begin{matrix}{{\overset{\sim}{C}}_{m} = {C_{m} - {\sum\limits_{k}{f_{k}^{+}S_{m - k}^{+}}} - {\sum\limits_{k}{f_{k}^{-}S_{m - k}^{-}}}}} & ( {{equation}\quad 1} )\end{matrix}$where:S_(m) ⁺ is the sample m of the upper satellite signal;S_(m) ⁻ is the sample m of the lower satellite signal;C_(m) is the sample m of the main signal;f_(k) ⁺ are the coefficients of the upper filter and f_(k) ⁻ are thecoefficients of the lower filter;{tilde over (C)}_(m) is the improved main sample m obtained at theoutput the subtractor 93.

Advantageously, the algorithm used to update the filter coefficients isthe LMS algorithm (Least Mean Square). According to the invention thedriving term Z_(m) of the algorithm (that is the term to be minimized)is the mismatch between the actual and the expected run length betweentwo transitions of the main signal.This means that: $\begin{matrix}{( f_{k}^{\pm} )_{m + 1} = {{( {1 - \mu} )( f_{k}^{\pm} )_{m}} - {\mu\frac{\partial}{\partial f_{k}^{\pm}}( Z_{m} )^{2}}}} & ( {{equation}\quad 2} )\end{matrix}$

The cross-talk minimization scheme of the invention will be describedbelow with reference to FIG. 6. As will be apparent from thisdescription, for the proposed scheme to operate properly, the ratio αbetween the PLL-driven bit clock and the fixed clock that runs theanalog-to-digital converters 51, 52 and 53 must be available. In FIG. 5,two arrows 141 and 143 indicate that the frequency ratio α is suppliedto the first and second coefficient updating means 111 and 113. Thearrows 141 and 143 are represented in dashed lines because they may beomitted if the frequency ratio α is equal to 1.

The ratio α is advantageously supplied by a time recovery circuitexternal to the cross-talk cancellation means 42 and specificallydesigned for fast approximate recovery of the bit frequency. Such anexternal time recovery circuit is already present in most readingapparatuses.

For example, in some systems (mostly in writable/rewritable systems),the wobble clock can be conveniently used for estimating the ratio α.FIG. 1 gives an example of an implementation that will be advantageouslyused in such systems: in FIG. 1, the external time recovery circuitcarries reference number 50-1 and is connected In the path between themain reading element 12 and the cross-talk cancellation means 42. Inother systems (mostly in ROM systems), average run length measurementscan be used for the same purpose. FIG. 2 gives an example of animplementation that will be advantageously used in such systems: in FIG.2, the external time recovery circuit carries reference number 50-2 andis connected In the path between the cross-talk cancellation means 42and the decoding means 44.

FIG. 5 is a schematic representation of the received main signal. Twosuccessive transitions X_(m) and X_(m+1), are represented.

{tilde over (C)}_((m,L)) is the improved main sample on the left of thetransition X_(m);

{tilde over (C)}_((m,R)) is the improved main sample on the right of thetransition X_(m);

{tilde over (C)}_((m+1,L)) is the improved main sample on the left ofthe transition X_(m+1);

{tilde over (C)}_((m+1,R)) is the improved main sample on the right ofthe transition X_(m+1);

{tilde over (C)}_((m,L)+1) is the improved main sample that precedessample {tilde over (C)}_((m,L));

{tilde over (C)}_((m,L)+1) is the improved main sample that followssample {tilde over (C)}_((m,L));

{tilde over (C)}_((m,R)−1) is the improved main sample that precedessample {tilde over (C)}_((m,R));

{tilde over (C)}_((m,R)+1) is the improved main sample that followssample {tilde over (C)}_((m,R));

φ_(m) is the time interval between the ideal time of the transitionX_(m) and the actual time of the transition X_(m) (in FIG. 5, φ_(m)<0);

φ_(m+1) is the time interval between the ideal time of the transitionX_(m+1) and the actual time of the transition X_(m+1) (in FIG. 5,φ_(m+1)<0);

d_(m+1,m) is the actual run length between the two transitions X_(m) andX_(m+1).

In the following it is assumed for simplification purposes, without lossof generality, that:

the time interval between two samples is equal to 1,

the transition moment m=1 corresponds to a rising transition,

and the time interval φ_(m) takes values from the interval$\lbrack {{- \frac{1}{2}},{+ \frac{1}{2}}} \rbrack.$

A first implementation of the updating scheme of the invention will nowbe described. This first implementation is applicable when the fixedsystem clock is (nearly) equal to the PLL driven bit clock (that is whenthe ratio α is close to 1), but there is no phase lock between the twoclocks.

The LMS driving parameter to be minimized Z_(m) is chosen to be equal tothe difference between the actual run length d_(m+1,m) and the expectedrun length d_(m+1,m) ^((exp)). Taking into account that an integernumber of clock intervals should ideally fit between the transitions inthe RLL encoded signal when there is no inter-symbol interference and noclock frequency variations, d_(m+1,m) ^((exp)) can be approximated asd_(m+1,m) ^((exp))=round(d_(m+1,m)) where round(x) is defined as theinteger number that is closest to the real number x.

Thus:Z _(m)=ζ(d _(m+1,m))where ζ(x)=x−round(x)with d _(m+1,m)=[(m+1, L)−(m,L)]+φ_(m+1)−φ_(m)  (equation 3)where [(m+1,L)−(m,L)] denotes the integer number of sampling intervalsbetween the samples {tilde over (C)}_((m,L)) and {tilde over(C)}_((m+1,L)).In the following it is assumed that the cross-talk is not extremelylarge, so that for small variations of the filter coefficients${{\zeta( d_{{m + 1},m} )}} < {\frac{1}{2}.}$With this assumption, Z_(m) can be approximated as follows:Z_(m)=ζ(d_(m+1,m))≈φ_(m+1)−φ_(m)+E where E is an integer independent ofthe filter coefficients. $\begin{matrix}{ \Rightarrow\frac{\partial Z_{m}}{\partial f_{k}^{\pm}}  = \frac{\partial( {\varphi_{m + 1} - \varphi_{m}} )}{\partial f_{k}^{\pm}}} & ( {{equation}\quad 4} )\end{matrix}$

The time interval φ_(m) can be computed approximately as a functiong_(m) of the improved main samples:φ_(m) ≈g _(m)({tilde over (C)} _((m,L)) ,{tilde over (C)} _((m,L)−1) , .. . ,{tilde over (C)} _((m,L)−N) _(L) ,{tilde over (C)} _((m,R)−1) , . .. ,{tilde over (C)} _((m,R)−N) _(R) )·(−1)^(m)The general form of a linear approximation is:$\varphi_{m} \approx {( {{\sum\limits_{k = 0}^{N_{L}}{\eta_{k,L}{\overset{\sim}{C}}_{{({m,L})} - k}}} + {\sum\limits_{k = 0}^{N_{R}}{\eta_{k,R}{\overset{\sim}{C}}_{{({m,R})} - k}}}} ) \cdot ( {- 1} )^{m}}$A simple 2-term linear approximation may be used, which gives:φ_(m)≈η·({tilde over (C)} _((m,L)) +{tilde over (C)} _((m,R)))·(−1)^(m)with η_(k,L)=η_(k,R)=η>0and φ_(m+1)≈η·({tilde over (C)} _((m+1,L)) +{tilde over (C)}_((m+1,R)))·(−1)^(m+1)Based on this simple 2-term linear approximation and on equation 3above:Z _(m)≈ζ(ηR _(m)·(−1)^(m+1)+[(m+1,L)−(m,L)])=ζ(ηR _(m))·(−1)^(m+1)where R _(m) ={tilde over (C)} _((m,L)) +{tilde over (C)} _((m,R))+{tilde over (C)} _((m+1,L)) +{tilde over (C)} _((m+1,R))The term $\frac{\partial}{\partial f_{k}^{\pm}}( Z_{m} )^{2}$in equation 2 can be computed as follows:${\frac{\partial}{\partial f_{k}^{\pm}}( Z_{m} )^{2}} =  {{2 \cdot Z_{m} \cdot \frac{\partial Z_{m}}{\partial f_{k}^{\pm}}} \approx {2 \cdot {\zeta( {\eta\quad R_{m}} )} \cdot ( {- 1} )^{m + 1} \cdot \frac{\partial( {\varphi_{m + 1} - \varphi_{m}} )}{\partial f_{k}^{\pm}}}}\Rightarrow{{\frac{\partial}{\partial f_{k}^{\pm}}( Z_{m} )^{2}} \approx {2 \cdot \eta \cdot {\zeta( {\eta\quad R_{m}} )} \cdot \lbrack {S_{{({m,L})} - k}^{\pm} + S_{{({m,R})} - k}^{\pm} + S_{{({{m + 1},L})} - k}^{\pm} + S_{{({{m + 1},R})} - k}^{\pm}} \rbrack}} $Eventually the expression for updating the filters coefficients is:(f _(k) ^(±))_(m+1)=(1−μ)(f _(k) ^(±))_(m)−2·μ·η·ζ(ηR _(m))·(S_((m,L)−k) ^(±) +S _((m,R)−k) ^(±) +S _((m+1,L)−k) ^(±) +S _((m+1,R)−k)^(±))  (equation 5)

A second implementation of the updating scheme of the invention will nowbe described that can be used when the fixed system clock (under whichthe filters are running) is not equal to the PLL driven bit clock (thatis when the ratio α≠1).

In this second implementation, the LMS driving parameter to be minimizedZ_(m) is also chosen to be equal to the difference between the actualrun length d_(m+1,m) and the expected run length d_(m+1,m) ^((exp)), butthe mathematical formulae used for computing d_(m+1,m),φ_(m) and φ_(m+1)have to be modified so as to take into account the frequency ratio α.

Namely, in order to measure the run length in bit intervals, the numberof samples between two transitions has to be multiplied by α, whichmeans that:d _(m+1,m)=α·[(m+1,L)−(m,L)]+φ_(m+1)−φ_(m)The transition phases φ_(m) also have to be multiplied by α. This meansthat the general form of the linear approximation of φ_(m) is:$\varphi_{m} \approx {\alpha \cdot ( {{\sum\limits_{k = 0}^{N_{L}}{\eta_{k,L}{\overset{\sim}{C}}_{{({m,L})} - k}}} + {\sum\limits_{k = 0}^{N_{R}}{\eta_{k,R}{\overset{\sim}{C}}_{{({m,R})} - k}}}} ) \cdot ( {- 1} )^{m}}$and the simple 2-term expression of the linear approximation is:

φ_(m)≈α·η·({tilde over (C)} _((m,L)) +{tilde over (C)}_((m,R)))·(−1)^(m)Eventually the expression for updating the filters coefficients is:(f _(k) ^(±))_(m+1)=(1−μ)(f _(k) ^(±))_(m)−2·α·μ·η·ζ(ηR _(m))·└S_((m,L)−k) ^(±) +S _((m,R)−k) ^(±) +S _((m+1,L)−k) ^(±) +S _((m+1,R)−k)^(±)┘  (equation 6)

It will be noted from equations 5 and 6 that the minimization scheme ofthe invention does not use the notion of ideal transition time.

With respect to the described cross-talk cancellation method, signalprocessor and reading apparatus, modifications or improvements may beproposed without departing from the scope of the invention. Theinvention is not limited to the examples provided. In particular:

The first and second implementations that were described are based on asimple 2-term linear approximation for the calculation of the timeintervals φ_(m) and φ_(m+1). This is not restrictive. Otherapproximations can be used. For example, a linear approximation usingmore than 2 terms may be used. The LMS updating scheme for these otherapproximations can be derived in a similar fashion as for the 2-termlinear approximation.

The minimization algorithm used in the above described implementationsis the LMS algorithm. This is not restrictive. Other minimizationalgorithms may be used to minimize Z_(m). The corresponding coefficientupdating equations may be readily derived by using the same principlesas those described above for the LMS algorithm.

In the cross-talk cancellation means described with reference to FIG. 5,the main signal is equalized. In an alternative embodiment, the mainsignal may be processed by an adaptive filter in a similar fashion asthe lower and upper satellite signal.

The functions described above may be implemented either in hardware orin software. FIGS. 1, 2 and 5 are functional representations of anapparatus and a signal processor according to the invention. A hardwareimplementation thereof may differ from this functional blockrepresentation.

The word “comprising” does not exclude the presence of elements or stepsother than those listed.

1. A cross-talk cancellation method using a main signal (62) associatedwith a target track (32) and satellite signals (61, 63) associated withside tracks (31, 33), said main signal showing transitions (X_(m)) andruns of various lengths (d_(m+1,m)) between two transitions(X_(m),X_(m+1)), said cancellation method comprising the steps of:filtering said satellite signals with adaptive filters (71, 73), therebygenerating filtered versions (81, 83) of said satellite signals,updating the coefficients of said adaptive filters by minimizing themismatch between the actual and the expected run length between twotransitions of the main signal, and processing said main signal, therebygenerating an improved main signal (102), said processing including asubtraction of said filtered versions of said satellite signals.
 2. Aprogram comprising instructions for implementing a cross-talkcancellation method as claimed in claim 1 when said program is executedby a processor.
 3. A signal processor (40) comprising cross-talkcancellation means (42) for receiving a main signal (62) associated witha target track (32) and satellite signals (61, 63) associated with sidetracks (31, 33), said main signal showing transitions (X_(m)) and runsof various lengths (d_(m+1,m)) between two transitions (X_(m),X_(m+1)),said cross-talk cancellation means comprising: filtering means (71, 73)for filtering said satellite signals with adaptive filters, therebygenerating filtered versions (81, 83) of said satellite signals,updating means (111, 113) for updating the coefficients of said adaptivefilters by minimizing the mismatch between the actual (d_(m+1,m)) andthe expected (d_(m+1,m) ^((exp))) run length between two transitions ofthe main signal, and processing means (93) for generating an improvedmain signal (102) from said main signal by subtraction of said filteredversions of the satellite signals.
 4. A signal processor as claimed inclaim 3, comprising a fixed clock (55), time recovery means (130), and abit clock (120) driven by said time recovery means, said fixed clockbeing asynchronous with respect to said bit clock, wherein saidcross-talk cancellation means are operated at said fixed clock.
 5. Asignal processor as claimed in claim 4, wherein said bit clock has a bitclock frequency and said fixed clock has a fixed clock frequency that issubstantially different from said bit clock frequency such that theratio between said bit clock frequency and said fixed clock frequency issubstantially different from 1, said signal processor further comprisingtime recovery means (50-1, 50-2) for estimating said ratio and providingsaid ratio to said updating means, said updating means being designed totake said ratio into account for updating said coefficients.
 6. Anapparatus (6-1, 6-2) for reading a signal stored along a track on astorage medium (1) comprising a signal processor as claimed in claim 3.7. An apparatus for reading a signal stored along a track on a storagemedium comprising a signal processor as claimed in claim
 4. 8. Anapparatus for reading a signal stored along a track on a storage mediumcomprising a signal processor as claimed in claim 5.